Data transmission using frequency shift keying (FSK) modulation of a radio frequency (RF) carrier wave is widely employed for transmitting digital data. A special case of spectrally efficient FSK is known as minimum shift keying (MSK). In MSK, two orthogonal signals represent the binary values 0 and 1. Typically a binary one is represented by a first frequency (f1) and a binary zero is represented by a second frequency which equals f2; the first and second frequencies have the same AC amplitude. Generators of MSK signals usually include an I-Q modulator having an input responsive to a binary data source and two mixers (that is, signal multipliers) responsive to orthogonal components of a carrier. The data rate of an MSK system is determined by the maximum frequency shift, i.e., frequency deviation, of the transmitted signal from the frequency of a carrier wave. To preserve the orthogonal nature of MSK, the peak to peak frequency deviation equals the bit data rate divided by 2. For example, a typical very low frequency (VLF; between 3 kHz and 30 kHz) system in MSK mode with a frequency deviation of +/−50 Hz (i.e. 100 Hz peak to peak deviation) has a data rate equal to 100×2=200 bits per second. Any increase or decrease in data rate of an MSK system requires a corresponding change in frequency deviation.
MSK is often used in systems having transmit antennas with restricted useful bandwidth (typically 1 dB or less) because MSK is spectrally efficient. The wavelength of the RF carrier frequency frequently used in the VLF band is typically in the range of 10 to 30 kilometers. It is impractical to build a transmitting antenna large enough to be a significant fraction of these wavelengths. The typical VLF antennas, e.g., the antennas at the stations operated by the United States Navy in Maine and Hawaii for underwater radio transmission to submarines, occupy about a thousand acres of land area and still are only a small fraction of a wavelength in height, despite having multiple transmitting towers that respectively have heights of 304 meters (997.5 feet) (about 0.03 wavelength at the lowest VLF frequency) and 458.1 meters (1503 feet). (about 0.046 wavelength at the lowest VLF frequency). The economics of land and construction costs put practical limits on the size of any high power VLF antenna; the tower in Maine radiates 1800 kilowatts of power at a frequency of 24.0 kHz, but cannot handle digital signals having a rate greater than 200 bits per second.
Based on the above, the useful bandwidth of a typical high power transmitter including a VLF antenna is much less than a typical transmitter having an antenna for higher frequency bands. The useful bandwidth of a typical transmitter including a VLF antenna is in the range of 25-100 Hz. The maximum data rate that can be transmitted by existing high power VLF transmitters is limited by the antenna system useful bandwidth of these transmitters.
One advantage of FSK and MSK is that the resulting RF signal has constant amplitude. Typical transmitter power levels for high power VLF transmitting stations are in the range of 100 kW to 2,000 kW. Therefore, high efficiency is a key requirement to minimize operational cost. Because the transmitted signal has a constant amplitude envelope it can be amplified by simple power amplifiers that operate in high efficiency modes, such as Class C or Class D. For this reason, all prior art high power VLF transmitters utilize these types of high efficiency amplifiers and are incapable of handling any other type of modulation such as AM.
FIG. 1 is a block diagram of a prior art high power VLF transmitter employing MSK modulation. The transmitter of FIG. 1 is responsive to binary data source 910 having an output which supplies a bi-level, non-return to zero (NRZ) signal to MSK generator 912 which is responsive to VLF carrier source 916 and derives a frequency coded output, i.e., a variable frequency output dependent on the binary values of the output of source 910. In response to source 910 deriving binary one and zero values, generator 912 respectively derives first and second frequencies having the same AC amplitude at the carrier frequency minus the deviation frequency, and the carrier frequency plus the deviation frequency.
The MSK output of generator 912 is supplied to transmitter 914. Transmitter 914 includes a high power, high efficiency amplifier, such as Class C vacuum tube amplifier 915 including a tuned circuit having a resonant frequency equal to the VLF carrier frequency, or Class D transistor amplifier 917 including a comparator for converting the RF input signal into a square wave, with provision to provide envelope modulation if necessary by changing the number of operating amplifiers on every RF cycle. Transmitter 914 also includes antenna impedance matching network 918, which is responsive to the output of the Class C or Class D amplifier, as appropriate.
If the data (or bit) rate of source 910 is relatively low, no greater than 200 bits per second in the installations in Maine and Hawaii, network 918, in turn, supplies an MSK signal having an envelope with constant amplitude to high power VLF electromagnetic wave antenna system 920. Under such circumstances, antenna system 920, such as the previously described antenna systems in Maine and Hawaii, emits a VLF band wave with modulation having a substantially constant amplitude envelope with modulation having a wave shape that is a substantial replica of the wave shape derived by MSK generator 912.
The total frequency response of the cascaded sub-elements of the transmitter system of FIG. 1 can be found by taking the convolution of the impulse response of each of the sub-elements. In the block diagram of FIG. 1, antenna system 920 and matching network 918 cause the transmitter system of FIG. 1, (as described to this point) to have an extremely narrow useful bandwidth. In the time domain, this narrow bandwidth causes errors in the transmitted waveform that increase rapidly with increasing data rate, particularly above 200 bits per second in the transmitters in Hawaii and Maine. The impulse responses of antenna system 920 and matching network 918 cause these errors in the time domain.
If the bit rate of source 910 is higher than a certain level, such as 200 bits per second, the components of transmitter 914 (as described to this point), matching network 918, and particularly antenna system 920 have frequency responses and group delay distortion (that is, an error in the relative time delay across the bandwidth of the antenna system 920 and the components ((as described to this point)), between the antenna system and the output of generator 912) that change the shape of the frequency modulated wave which MSK generator 912 derives so that the shape of the modulated wave emitted by antenna system 920 is not a replica of the wave that generator 912 derives. Transmitter 914 (as described to this point), matching network 918 and particularly antenna system 920 cannot accurately replicate the sidebands, especially the higher order sidebands, associated with accurate reproduction of the higher bit rate frequency modulated wave derived by MSK generator 912. (The reader will recall that a frequency modulated wave is theoretically represented by an infinite number of higher order terms having coefficients represented by Bessel functions.) Because the modulation wave emitted by antenna system 920 is not an accurate replica of the wave derived by generator 912, under these circumstances, the signal at a receiver responsive to the wave emitted by antenna system 920 does not accurately replicate the output of binary data source 910.
Systems of the type described above have the disadvantages noted above relating to low data rate and are massive, highly expensive structures occupying enormous areas. In addition, considerable stresses are exerted on antenna system 920 in response to transients in the modulated wave that matching network 918 supplies to the antenna system. For example, discharges sometimes occur across insulators of the antenna system, which insulators maintain components of the antenna system ungrounded.
Many of these disadvantages are overcome by the transmitter system disclosed in the commonly assigned U.S. Pat. No. 8,355,460 which, as illustrated in FIG. 1 hereof, includes arbitrary impulse response pre-corrector 922, comparator 924, least mean square (LMS) calculator 926, real component current detector 928 and delay element 930. Another difference between the prior art described above and the transmitter system of the '460 patent is that the MSK generator of the '460 patent is not responsive to the carrier source. Instead, in the '460 patent, the MSK generator derives a frequency coded signal at baseband and the transmitter is modified so it responds to a baseband output of pre-corrector 922 and carrier source 916 to derive a VLF or LF carrier with frequency and amplitude modulation. The modulation is an amplified replica of the shape of the input signal pre-corrector 922 supplies to transmitter 914. The construction and operation of elements 922, 924, 926 and 930 are described in detail in the '460 patent and need not be described herein, except to note that they cause the output of transmitter 914 to be amplitude modulated so the power radiated from antenna 920 is relatively constant for data rates of source 910 up to 600 bits per second.
While discrete components are illustrated in FIG. 1 for components 912, 914, 916, 922, 924, 926, and 930 for convenience and ease of explanation, it is to be understood that many or all of the operations performed by these discrete components, except those related to power amplification and power handling, can be and are preferably performed numerically, in a computer. For, example, carrier source 916 can derive the carrier as a series of relatively small steps by using a sum of Walsh functions approximating a modulated sine wave. If a Walsh function sum generator is employed, the steps are smoothed by a low pass filter (not shown), having a cut off frequency of about 1 megahertz, and that drives antenna matching network 918.
FIGS. 2 and 3 are equivalent circuit diagrams of the distributed impedances of typical VLF antenna systems, such as antenna system 920 or the antenna systems included in the transmitter systems described infra in connection with the detailed description of the present invention. Such antenna systems typically include two or three towers, each having a height of about 500 meters, with many umbrella wires. The antenna system is typically driven by a transmitter system having (1) an output of about 500 kW, and (2) radiation efficiency in the range of about 65 to 80 percent. The antenna systems have electrical lengths, at VLF, of about 0.03-0.048 wavelengths with a low radiation resistance and a high series capacitive reactance. Resonating the capacitance with a helix inductor, causes the transmitter to produce a voltage up to 250 kilovolts. Because the antenna system is electrically very short, it has a high reactance, making impedance matching of the antenna to the transmitter difficult and generally possible over only a relatively narrow bandwidth of 0.2 KHz, except that the transmitter disclosed in the '846 patent has a bandwidth of about 0.6 KHz.
The circuit diagram of FIG. 2 assumes that the equivalent circuit of antenna system 920 is a series circuit including 417.99 microhenry inductor 931, which represents the inductance of the antenna system, 72519.3 picofarad capacitor 932, which represents the capacitance of the antenna system, and 0.1030 ohm resistor 933, which represents the radiation resistance of antenna system 920 at 16 kHz, the carrier frequency of transmitter 914 associated with the matching networks of FIGS. 7, 10, 12 and 14. The antenna system 920 represented by the equivalent diagram of FIG. 2 also includes 440 picofarad shunt capacitor 937, which represents the capacitance of a bushing of antenna system 920, as represented by FIG. 2, and the series combination of 0.05 ohm resistor 938, which represents the resistance loss of the earth in the vicinity of the antenna system, and 0.03505 ohm resistor 939 which represents the antenna system loss due to factors other than those represented by resistors 933 and 938.
The foregoing component values are associated with the antenna systems of FIGS. 7, 10 and 12. The component values of the different antenna systems associated with the matching networks of FIGS. 14, 18, 20 and 22, as represented by the diagram of FIG. 2, are discussed infra.
The equivalent circuit diagram of antenna system 920 illustrated in FIG. 3, includes 167.5 microhenry inductor 934, which is connected in series with the parallel combination of 92400 picofarad capacitor 935 and 62000 ohm resistor 936. The 62000 value of resistor 936 represents the parallel-equivalent radiation resistance at 12 kHz of antenna system 920 in the parallel configuration of FIG. 3. The antenna system 920 represented by the equivalent diagram of FIG. 3 also includes 530 picofarad shunt capacitor 937, which represents the capacitance of a bushing of antenna system 920 as represented by FIG. 3, and the series combination of 0.077 ohm resistor 938, which represents the resistance loss of the earth in the vicinity of the antenna system, and 0.0652 ohm resistor 939 which represents the antenna system loss due to factors other than those represented by resistors 933 and 938. The antenna system of FIG. 3 is associated with the matching network of FIG. 16.
The radiation resistance of the prior art narrow band antenna systems can be and has been assumed to be constant as a function of frequency; such an assumption cannot be made with the wider band transmitter systems described supra in connection with the detailed description of the present invention.
FIG. 4 is a circuit diagram of a typical prior art impedance matching network of a typical VLF transmitter system, such as network 918 of FIG. 1. The voltage output of a Class D, high power amplifier 917 of transmitter 914 drives terminal 940 of the matching network. The voltage at terminal 940 is supplied to primary circuit 961 of the network, which primary circuit contains cascaded phase shifters 941 and 942, each of which is (1) a low pass filter for removing unwanted harmonics, and (2) a 90 degree phase shifter at the carrier frequency of the output of transmitter 914. Each of phase shifters 941 and 942 has a tee configuration so that phase shifter 941 includes series inductors 943 and half of inductor 944 and shunt capacitor 945, while phase shifter 942 includes the series combination of the other half of inductor 944 and inductor 946, as well as shunt capacitor 947.
The phase shifted voltage at terminal 948 of phase shifter 942 is applied to tap 949 of autotransformer 950. Autotransformer 950 includes variable inductor 951 between tap 949 and terminal 952, as well as fixed inductor 953 between tap 949 and terminal 954. The voltage at terminal 949 is applied to antenna system 920 by the series combination of fixed inductor 953, variable inductor 955 and fixed inductor 956. The inductance of inductor 955 is adjusted such that the series combination of inductors 953, 955 and 956 and the antenna system impedance, as represented by the components of either FIG. 2 or 3, is resonant to the carrier frequency of the output of transmitter 914. The resonant condition is a factor in causing the bandwidth of the transmitter system including the network of FIG. 4 to be narrow. Inductors 953, 955 and 956, as well as antenna 920, form a secondary circuit 963 of the matching network. Primary circuit 961 and secondary circuit 963 are coupled to each other by variable inductor 951, which provides shunt inductive coupling between the primary and secondary circuits 961 and 963.
FIG. 5 includes a response curve 957 of voltage amplitude (in dB) vs. frequency of a typical prior art impedance matching network, such as the network illustrated in FIG. 4. Response curve 957 includes two peak values, at the tops of horns 958 and 959 that are approximately equidistant from a center frequency (the transmitter carrier frequency). Response curve 957 closely resembles the response curve (voltage versus frequency response) of an overcoupled doubly tuned filter, i.e., a filter having two resonances at the frequencies of the horns. Response curve 957 has a center frequency of 21.0 kHz and horns 958 and 959, at about 20.69 kHz and 21.32 kHz, which horns have peak voltages about 22.5 dB above the center frequency voltage; the center frequency corresponds with the carrier frequency of transmitter 914. Thus the frequency difference between horns 958 and 959 is about 600 Hz and the ratio of the frequency difference of the horns to the carrier frequency is about 0.0286. The transmitter system of FIG. 1, in combination with the matching network of FIG. 4 is able to handle bit rates of source 910 up to about 300 bits per second that are coded in minimum shift keying or Gaussian minimum shift keying.
Phase shifters 941 and 942, in combination with inductors 951, 953, 955 and 956, form a reactive impedance matching network between the output of transmitter 914, at terminal 940, and antenna system 920. In addition, phase shifters 941 and 942 are a low pass filter that suppresses harmonics of the carrier frequency introduced by stepped sine wave formation of the carrier or a square wave carrier.
Variable inductor 951 provides shunt inductive coupling between (1) the output (at terminal 948) of primary circuit 961 including phase shifters 941 and 942, and (2) secondary circuit 963 including inductors 953, 955 and 956, as well as the impedances (resistance, inductance and capacitance) of antenna system 920. The coupling provided by inductor 951 is determined by the necessary impedance transformation between the output of transmitter 914, at terminal 940, and antenna system 920. Consequently, the frequency separation between horns 958 and 959 is essentially fixed and determined by the reactances of antenna system 920 and the impedance looking into phase shifter 941 that is presented to the output of transmitter 914. If it is desired to change the coupling between terminal 948 at the output of phase shifter 942 and antenna system 920 and thereby change the spacing between horns 958 and 959, the inductance of variable inductor 951 is changed. However, the change in inductance of inductor 951 must be accompanied by a change in the inductance of inductor 955 to maintain the center frequency of response 957 between horns 958 and 959 at the carrier frequency of transmitter 914. Changing the inductance of inductor 955 to maintain the same impedance detunes the matching network of FIG. 4. Therefore, coupling and bandwidth in the matching network are essentially fixed.
It is, accordingly, an object of the present invention to provide a new and improved transmitting system, particularly adapted to operate in the VLF range or the low-frequency (LF; from 30 kHz to 300 kHz) range, wherein the transmitter system has a relatively wide bandwidth.
Another object of the invention is to provide a new and improved transmitting system, particularly adapted to operate in the VLF range or the LF range, wherein a matching network of the transmitter system can be easily adjusted to efficiently handle different bandwidth signals.
A further object of the invention is to provide a new and improved transmitting system, particularly adapted to operate at LF or VLF, wherein a matching network of the transmitter system can be adjusted with a single step to efficiently handle different bandwidths.
An additional object of the invention is to provide a new and improved transmitting system, particularly adapted to operate at LF or VLF, wherein a matching network of the transmitter system has a voltage amplitude versus frequency response that resembles the response of an overcoupled doubly tuned network because it includes a pair of horns, and the spacing between the horns can be adjusted independently of the impedance transformation the matching network provides.
An added object of the invention is to provide a new and improved transmitting system, particularly adapted to operate at LF or VLF, wherein a matching network of the transmitter system enables such a transmitter system to have a greater bandwidth than prior art transmitter systems, and coupling between primary and secondary circuits of the matching network is easily adjusted and the coupling is independent of the matching network impedance ratio, that is, the ratio presented to the matching network input terminals by the output of the transmitter power amplifier to the impedance of the antenna system the matching network drives.
A still further object of the invention is to provide a method of modifying existing transmitting systems adapted to operate in the VLF or LF ranges, wherein coupling between primary and secondary circuits of a matching network of the transmitter system is modified to enable such transmitter systems to: (1) have a greater bandwidth, (2) be easily adjusted, and (3) be adjusted independently of the matching network impedance ratio, that is, the ratio of (a) the impedance presented to the matching network input terminals by the output of the transmitter power amplifier to (b) the impedance of the antenna system the matching network drives.